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14- Digital modulation techniques

14- Digital modulation techniques - Digital modulation...

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Unformatted text preview: Digital modulation techniques for mobile and personal communication systems This paper describes in detail various digital modulation techniques for mobile and personal communication systems. Among others, these include 77." /4-shift quadrature phase-shift keying (QPSK), used in the second generation digital cellular mobile systems in North America and Japan, and Gaussian minimum-shift keying (GMSK), employed in the GSM system in Europe. It then briefly discusses the current research activities in modulation schemes for future systems. by A. H. Aghvami 1 Introduction 0 constant or near constant The choice of modulation envelope technique has a direct impact on 0 low cost and ease of the capacity of a digital cellular implementation, mobile (DCM) communication system. It determines the bandwidth efficiency of a single To optimise all these features at the same time is not possible as each physical channel in terms of the has its practical limitation and also number ofbits per second per is related to the others. For hertz (bil/s/Hz) and it is therefore example, to achieve high important that this choice is bandw1dth efficiency one may discussed in detail. choose to use hlgh-level modulation. However, if this is done two consequent disadvantages are introduced. Firstly, the power efficiency of the system is reduced. Secondly, the In selecting a suitable modulation scheme for a DCM system, consideration must be given to achieving the following: 0 hggh bandWIdth éfficwncy bandlimited high-level modulated . hlgh power efflCIency signal has a large envelope 0 low carrier-to»cochannel variation which, when the signal is interference power ratio (CCI) passed through a power efficient 0 low out-of—band radiation nonlinear amplifier, generates 0 low sensitivity to multipath large out—of-band radiation; this, in fading turn, introduces interference to inphase branch U/B converter incoming bit stream U/B converter quadrature branch 1 QPSK modulator ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 adjacent channels. The latter disadvantage can be circumvented by using linear power amplifiers but these have poor power efficiency. Hence, a trade-off among all the above features must be adopted. In this paper, the power efficient QPSK, OK-QPSK, Ir/4-shift QPSK and GMSK modulation techniques are first described and then the modulation techniques for future DCM systems are discussed. 2 Quadrature phase shift keying (QPSK) systems Quadrature PSK (also referred to as 4-ary PSK or 4-phase PSK) modems (modulator/ demodulators) are extensively employed in operational satellite communication systems. The block diagram of a conventional QPSK modulator is shown in Fig. l. The input unipolar binary stream at a bit rate of fh is converted into two bit streams (inphase and quadrature streams), each having a bit rate of j: =fb/2 (the symbol rate) by a serial-to- parallel (S/P) converter. The two unipolar-to-bipolar (U/B) converters convert these two streams into two bipolar (i 1) binary signals, which are passed through two spectrally shaped lowpass filters and then modulated by the inphase and quadrature carriers. The modulation operation uses the double-sideband suppressed—carrier amplitude bandlimited QPSK signal 125 incoming bit stream inphase bit stream quadrature bit stream 2 Operation of serial-to-parallel converter modulation (DSS-SC—AM) technique. The two modulated signals (each of which can be considered as a binary PSK signal) are combined to give a QPSK signal. Finally, the QPSK signal is filtered at the output of the modulator to limit further its power spectrum; this prevents spill-over into adjacent channels and also removes out—of-band spurious signals caused by the modulation operations. It should be noted that the spectrally shaped filters can also be implemented as a bandpass filter at the output of the modulator. In this case, the lowpass filters shown in the block diagram are not required. quadrature T Fig. 2 illustrates the operation of the S/P converter. In QPSK systems, the modulated signal has four distinct phases (Ir/4, 371/4, 57r/4 and 77r/4), each representing one symbol. Each symbol contains 2 bits of information, The mapping of the bits into symbols is frequently done in accordance with the Gray code. This code ensures that a single symbol in error corresponds to a single bit in error. The constellation (signal—space diagram) of a Gray—coded unfiltered QPSK signal is shown in Fig. 3. This Figure shows that the carrier phase during any symbol interval can be any one of the four 3 Constellation of Gray-coded QPSK signal 126 phases 7r/4, 37r/4, 5717/4 and 77r/4 depending on the values of the inphase and quadrature components during that interval. Fig. 4 gives the block diagram of a coherent QPSK demodulator. The input bandpass filter removes the out-of—band noise and adjacent channel interference. The signal at the output of the filter is split into two parts, each part being coherently demodulated with the inphase and quadrature carriers. The two outputs are lowpass filtered and applied to one—bit analogue-to-digital (A/D) converters to regenerate the inphase and quadrature baseband signals. These two streams are finally recombined in a parallel—to- serial (P/S) converter to give the original bit stream. The carrier- recovery circuit provides inphase and quadrature carriers which are in synchronism with the received unmodulated signal, i.e. which have the same frequency and phase as the unmodulated inphase and quadrature carriers. The symbol timing recovery circuit provides the timing required to operate the two A/D converters, which must be in synchronism with the received baseband signals. Differential encoding/decoding for QPSK Most practical carrier recovery circuits introduce a phase ambiguity into the recovered carrier. In the case of QPSK, a four—phase ambiguity may occur, causing a considerable bit error rate.I To remove this phase ambiguity, a differential encoder may be employed in the modulator and a differential decoder in the demodulator. Differential encoding, ensures that the Changes in the phases of the transmitted carrier represent the information bits. Table 1 shows the phase shifts (phase advances) required for a QPSK signal to transmit all possible pairs of the information bits. The encoder can be implemented using logic circuits or look-up tables. The operation of a QPSK differential encoder is given in Table 2, where ak and bk Table 1 Phase shifts in differential encoding Information bits Phase shift 0 O O O 1 71/2 1 1 7c 1 O 371/2 ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 carrier recovery circuit received QPSK signal 4 Coherent QPSK demodulator are the kth uncoded inphase and quadrature symbols, respectively, at the output of the S/P converter, Ik and Qt represent the kth differentially encoded symbols, and [H and Q,“l are the kth differentially encoded symbols delayed by the symbol duration (Ts = 2Tb)~ In the case of differentially encoded QPSK (DEQPSK), the decoding operation is the inverse function of the encoding. For differential QPSK (DQPSK), the decoding operation is performed on the received modulated signal, thus avoiding the use of a carrier recovery circuit. Fig. 5 shows a differential QPSK demodulator block diagram. Spectrum and spectral efficiency of QPSK signals The power spectral density of an unfiltered QPSK is of the form: _ sin 27r(f—fc)Th 2 3‘“ ‘ 4C“ i 276— ffirfi “) one-bit A/D converter recovery symbol I timing P/S converter recovered bit stream oneebit circuit I A/ D converter where C is the average signal power normalised across a 1 Q resistor and Th = l/fb is the bit duration. Table 2 Differential encoder operation for QPSK “k bk Ik—l Qk—I [k Qk 0 0 0 0 0 O 0 0 0 1 0 1 0 0 l l l 1 0 0 1 0 1 0 0 1 0 O l 0 0 1 0 1 0 0 0 I l l O l 0 1 1 0 1 l l 1 0 O 1 l l 1 0 l l 0 1 l l l O 0 1 1 l O O 1 1 O O 0 0 l 1 0 0 1 l 1 l 0 l 1 l 0 I 0 1 0 O 0 Assuming that the modulator employs a spectrally shaped filter having a square root of a raised- cosine function with a roll-off factor of a (for optimum performance), the filtered spectrum ofa QPSK signal can easily be obtained, and is as shown in Fig. 6. The spectral efficiency of a modulation scheme is defined as the ratio of the bit rate to the bandwidth, fi/B, expressed in bit/s/Hz. In the case onPSK, the bandwidth of the signal is (l + 00f“ as in Fig. 6. Hence, its spectral efficiency is: fr )2 Tiorsx=E= (1+ mfs = since fl = fl/Z. For the minimum bandwidth case (a = 0), the theoretical spectral efficiency of QPSK systems is 2 bit/s/I-Iz. It is now possible to realise practical filters with a roll-off factor as low as 27 1+0: (2) a — 0-2 and hence a spectral onevbit A/D converter P/S converter one—bit A/D converter 5 Block diagram of QPSK differential demodulator ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 127 frequency 6 Power spectral densities of unfiltered and filtered QPSK signal: (a) unfiltered QPSK spectrum; (1;) amplitude response of raised-cosine function with roll-off factor (x and amplitude equaliser; (c) filtered QPSK spectrum nonlinear m ‘D 2‘ (I) C O) ‘c n: .. .s Q <1> a. (II ._ a: 3 o a linear 7 Spectrum spreading of bandlimited QPSK signal at output of TWI‘A operating at saturation4 efficiency of 17 bit/s/Hz can be achieved in practice. Error probability performance of QPSK systems The performance of a QPSK system in the presence of system impairments may be measured by its symbol error probability (symbol error rate) or bit error probability (bit error rate). The symbol error probability of a system is defined as the average probability of signalling elements of the received signal being in error (phase error in the case of PSK systems). The bit error probability of a system is defined as the average probability of the received bits being in error. In the presence of additive white Gaussian noise (AWGN) only and for transmission free of intersymbol interference (181), the expression for the symbol error probability of a QPSK system is: P5 = erfc [‘5 (%) J % (3) where C/N is the carrier-to-noise power ratio (CNR) and erfc(-) is the complementary error function given by: erfc(z) = 1 — erf(z) =fzje‘2dx , A. In practical measurements, it is convenient to measure the CN R. However, the bit energy (E b) to noise power spectral density (N0) ratio (E ,/N0) is a very useful parameter when various modulation schemes are compared. Assuming that the filtering strategy of the system satisfies the Nyquist criterion, 3 general relation between these two parameters for M-ary modulation schemes is given by: Q _ 1i (N) w (M) In the case onPSK, M = 4 and C _ a (r) -2 (Na) <5) By putting eqn.5 into eqn.3 the symbol error rate of a QPSK system as a function of E b/No is obtained as: PE = erfc (7)15 (6) Relationship oflhe bit error rate, PB, and symbol error rate, PE For Gray-coded M-ary modulation schemes the relationship between these two figures of merit can be easily 128 ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 unipolar to bipolar converter S/P Converter unipolar to bipolar converter derived as: 1 p = _ P 7 e logzM E ( ) In the case of the QPSK system we have: N (8) o a PE = app = %erfc (Eh) The BER performance of QPSK signals through nonlinear channels depends on the type of filtering and the amount of power amplification. A detailed treatment of the subject is given in Reference 2. 3 Offset-keyed QPSK (OK-QPSK) Most digital radio transmission systems today operate their main high power amplifiers (HPAs) near to or at saturation for maximum power efficiency. For this operating mode, the HPA introduces nonlinear amplitude and phase distortions. One of the damaging effects of these nonlinearities is the spectral spreading of the transmitted signal, which increases undesirable interference to the adjacent channels (adjacent channel interference). Because of their large phase transitions (i 90°, i 180°) QPSK signals, when bandlimited, have large envelope variations (up to 100%). As a result, when a bandlimited QPSK signal passes through a nonlinear amplifier operating at saturation. there is almost a total regeneration of the filtered sidelobes to their unfiltered levels (nearly 100% spectral spreading). Fig. 7 illustrates the spreading of a bandlimited QPSK oscillator r ----- 1 : sinusoidal 8 OK-QPSK and MSK modulator (--- for MSK only) quadrature inphase 9 Signal constellation of 7r/4-QPSK system signal by a nonlinear amplifier operating at saturation. The channel roll-off is assumed to be 40% (a = 0-4).3 For nonlinear systems, alternative modulation schemes with smaller phase transitions have been proposed. OK-QPSK signals have i 90° phase transitions. As a result, bandlimited OK-QPSK signals have a 33% envelope variation and hence. when they are transmitted through a nonlinear HPA, there is input bit stream mapping S/P converter 10 7r/4-shift QPSK modulator ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 bandlimited signal less spectral spreading than in the case of a bandlimited QPSK signal.4 A block diagram of an OK-QPSK modulator is shown in Fig. 8. The modulator is identical to the QPSK one except that the quadrature stream is delayed with respect to that in the inphase branch by one bit interval. The OK-QPSK demodulator is also identical to the QPSK one but with the inphase branch shifted with respect to the quadrature stream by T 1, seconds just before the P/S converter (see Fig. 4). For linear systems, the bit error rate and power spectral density of the OK-QPSK system are the same as those of QPSK systems. This modulation scheme has been proposed for aeronautical and land mobile satellite communication systems. 4 7r/4-shift QPSK fl/4-shift QPSK was first introduced by Baker in 1962.5 It is a compromise between QPSK and offset-keyed QPSK (OK-QPSK) in the sense that it has a maximum phase change of 135° compared to bandlimited Ir/4»shift QPSK 129 11 Baseband differential detector 1800 for QPSK and 90° for OK- QPSK. Hence, bandlimited 7r/4—shift QPSK has less envelope fluctuation than bandlimited QPSK but more than that of OK- QPSK. The main advantage of 7r/4-shift QPSK is that it can be noncoherently detected (by a differential detector or by an FM discriminator), which is not the case for OK—QPSK. Also, in the presence of multipath fading, it outperforms OK-QPSK. For these reasons. it has been selected for the Japanese and American second generation cellular digital mobile radio systems. In a n/4—QPSK modulator, signalling elements of the modulated signal are selected in turn from two QPSK constellations which are shifted by 71/4 with 12 IF differential detector 130 respect to each other, as shown in Fig. 9. The modulator block diagram is shown in Fig. 10. The input bit stream is converted by a serial-to- parallcl (S/P) converter into two parallel streams (at, bk). each with a symbol rate equal to half that of the incoming bit rate. The kth inphase and quadrature pulses (It and Q) at the output of the signal mapping circuit are determined by their previous pulse levels, 1H and QM, and the input symbols ak and bk as given by [k : IHcos 0k — OH sin 9;. Q = 1k_lsin 6k + QM cos 9k ak and bk, in turn, are related to the phase shift changes of the modulated signal according to converter converter P/S Table 3 Phase shifts as a function of information bits for 7r/4-shift QPSK Information bits Phase shift 0 0 It /4 0 1 3 7r/4 1 1 5 75/4 I 0 717/4 Table 3. The rest of the modulator is identical to that of a QPSK modulator. To demodulate the 7r/4-shift QPSK. we may use one of the following differential detection methods: (a) Basebanddifferential detection: In this method, the differential decoding is performed on the recovered inphase and quadrature baseband signals as shown in Fig. 11. It requires local oscillators but, since the phase error is removed by differential detection at baseband. phase coherence is not needed. (17) IF differential detection: Fig. 12 shows the block diagram of an 1F differential detector for fl/4-shift QPSK. Differential decoding is performed on the received IF signal using a delay line and two mixers and hence no local oscillator is needed in the receiver. (c) Limiter FM discriminator detection: The block diagram ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 l3 Limiter FM discrimator detector of an FM discriminator detector is shown in Fig. 13. The FM discriminator extracts the instantaneous frequency deviation of the received signal. The integrate- and—dump circuit integrates the frequency deviation over each symbol duration, the integral being the phase difference between two sampling instants. Finally, the 4-level threshold comparator detects the output phase difference. It has been shown that these schemes have practically the same performance in the presence of cochannel interference and Gaussian noise if the circuit elements are properly designed" The modulation adopted for the American digital cellular mobile (DCM) communication system (1354) is 7r/4-shift QPSK with a roll—off factor of (1:035. The burst transmit signal with a bit rate of 486 kbit/s through a 30 kHz channel bandwidth gives a spectral efficiency of 1-62 bit/s/HZ for this system. For the Japanese DCM system a 7r/4-shift QPSK modulation with a roll—off factor of (1:05 is used. The burst transmission bit rate and the physical channel bandwidth are 42 kbit/s and 25 kHz, respectively. This gives a bandwidth efficiency of 168 bit/S/Hz. 5 Gaussian minimum shift keying (GMSK) Minimum shift keying (MSK) is a special case of binary continuous- phase frequency shift keying (FSK) with a modulation index of 05 or with a frequency deviation of f2 — f1 = 1 2 4T,, where f2 and f1 are two discrete frequencies of the MSK signal representing logic states 1 and 0, Af= integrate and dump FM discriminator respectively, and Tb is the bit duration. The MSK modulator can be implemented using a voltage- controlled oscillator (VCO)l or a quadrature form similar to that of OK—QPSK. In the latter case, the rectangular shaped pulses before the two multipliers are converted to half-sinusoidal waveforms by two sinusoidal pulse shaped filters as shown in Fig. 8. The demodulator may also use coherent detection similar to that of OK—QPSK. For linear systems, the bit error rates of QPSK, OK-QPSK and coherent MSK are identical. An unfiltered MSK signal has a constant envelope and also a faster spectral roll-off than QPSK or OK-QPSK but its main-lobe OPSK/OK-OPSK I P/S - converter bandwidth is 15 times wider (see Fig. 14). For higher bandwidth efficiency and sharper cut-off, an MSK signal should be filtered before transmission. This introduces envelope variations of the bandlimited signal and as a result, when it is passed through a nonlinear high-power amplifier, its filtered power spectral density spreads. This may cause undesirable interference to the adjacent channels (adjacent channel interference). However, if a Gaussian-shaped filter is used for filtering the transmitted signal, the constant envelope property of the signal can be kept, since a premodulation lowpass filter (LPF) with Gaussian-shaped transfer fc+2f5 3 fc+2f5 frequency 14 Unfiltered power spectral densities of QPSK, OK-QPSK and MSK signals ELECTRONICS & COMMUNICATION ENGINEERING JOURNAL JUNE 1993 131 15 GMSK modulator 16 I’LL-TYPE GMSK modulator function is easily realised. A simple method for the modulator is to use a VCO directly with a premodulation Gaussian-shaped LPF as shown in Fig. 15. The impulse response of the filter is given by: h(t) = e‘lZ/ZUZTZ/O'TWZW) where 0' = \l(ln 2)/27rBT andB is the 3 dB bandwidth of the filter and T the bit duration. However, it is difficult to maintain the centre frequency accurately in an acceptable range. To overcome this drawback, a PLL-type GMSK modulator can be used. The block diagram of this modulator is shown in Fig. 16.7 The coherent demodulation t...
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