EEE445_Lectr29_applicatios

EEE445_Lectr29_applicatios - • Antenna systems Microwave...

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Unformatted text preview: • Antenna systems Microwave Engineering Applications To feed antenna by coplanar waveguide (CPW) Low insertion loss – TEM line Coupler for via‐less transformer – Collimation antenna system • GHz Electronic packaging – Coupled multi‐conductor multi‐layer transmission lines (MMTL) – MoM and FEM • Low‐loss wideband transmission – Integrated waveguides – FWG EEE 591/445 Lecture 28 Lecture 27 1 60GHz antenna system Japan Unlicensed No specified Europe US Unlicensed F ( GHz ) 55 57 59 61 63 65 67 Substrate Configuration for On-chip Antenna Air Metal1 Metal 6 r S i0 2 3 .9 Poly r 11 . 9 ~ 1000 s / cm 10 s / cm Si Lecture 27 2 60 GHz scanning antenna system • Rapid growth in wireless data link led industry towards utilizing the 57‐64 GHz unlicensed frequency band. – – – – high data rate GB/s. both LOS and non‐LOS. Suppress multi‐path fading. 3600 degree scanning capabilities. • Antenna array consists of wide angle sub‐arrays – Forming panoramic field of view. – well suitable to automotive and mobile communication needs at 77GHz. Lecture 28 EEE 591/445 Lecture 27 Research Plan 1. Architecture and operation. Various Phased-Array Topologies Based on MEMS TTD Devices: Modified Traveling Wave PA Chip T T T MEMS Chip Corporate Feed UWB Stable Beam PA Chip PA + SW Chip T T T T MEMS Chip MEMS Chip T T T T Half-Duplex Corporate Feed W/ TR switches TX RX Lecture 27 4 2. Antenna elements and feed structure: Slot-line Feed Network, Various Transitions, and RF coupling Vias Slotline (Antenna Lines) Duroid Substrates S W Finite Width Slotline Top/Bottom Grounds Slotline (MEMS Chip) g/4 TAS and feeding ms/ 4 Strip (RF Feed Line) slot/ 4 Slotline (MEMS Chip) Lecture 27 5 Model for inter-digital dipole antenna Port1 3D View a1 l1 Top View Unit: um t3 a2 Reflector h3 Gmax 2.28 0 -30 2.00 1.50 -60 60 1.00 0.50 -90 90 30 t1 s1 w1 s 2 s3 w2 t2 l1 477.0 h3 275 0o o 90 -120 120 -150 -180 150 Radiation patterns at 0 o 90o Lecture 27 Ansoft Corporation 0.00 XY Plot 1 Final Version3 Curve Info dB(S(LumpPort1,LumpPort1)) Setup1 : Sweep1 dB(S(LumpPort1, LumpPort2)) Setup1 : Sweep1 -5 -5.00 -10.00 S11 and S21 cur ves -15.00 Y1 -20.00 -25.00 -30.00 -35.00 -40.00 55.00 57.50 60.00 Freq [GHz] 62.50 65.00 Lecture 27 Initial design one No SiO2 layer transition SiO layer transition Top view 3-D view s w w 50um s 12.5um Lecture 27 Wider trace used here S11 and S21 Ansoft Corporation 0.00 XY Plot 1 m1 CPWA2CPS2 -5.00 Name X 60.0000 60.0000 Y -2.0731 -13.9035 Curve Info dB(S(LumpPort1,LumpPort1)) Setup1 : Sw eep1 dB(S(LumpPort1,LumpPort2)) Setup1 : Sw eep1 Sw m1 m2 Y1 -10.00 m2 -15.00 57.00 Freq [GHz] 62.00 Lecture 27 Initial design two Wider trace (W=50um S=12.5um) used here Top view 3-D view Lecture 27 S11 and S21 Ansoft Corporation 0.00 XY Plot 1 m1 CPWA2CPS3 -2.00 -4.00 Name X 60.0000 60.0000 Y -1.1457 -10.7620 Curve Info dB(S(LumpPort1,LumpPort1)) Setup1 : Sw eep1 dB(S(LumpPort1,LumpPort2)) Setup1 : Sw eep1 -6.00 Y1 -8.00 -10.00 -12.00 -14.00 57.00 m1 m2 m2 Freq [GHz] 62.00 Lecture 27 Optimized design Top view 3-D view Lecture 27 Configurations CPW connected to folded waveguide 1. Identical size with CPW from folded waveguide 2. Air or low dielectric substrate CPW to CPS transition 1. Trace width transition 2. SiO2 layer thickness transition 3. Wide band open band open CPS connected to dipole antenna dip 1. Identical substrate configuration with dipole antenna 2. Trace width is reduced for better impedance match Lecture 27 Size details CPW connected to folded waveguide CPS connected to dipole antenna s1 w1 l1 w1 150 s1 50 l1 400 Top view s2 w2 l2 w2 30 s2 12.5 l2 300 Side view CPW to CPS transition to CPS transition l3 r1 h1 w3 w3 150 h1 20 Lecture 27 Unit: um l3 80 r1 102 S11 and S21 Ansoft Corporation 0.00 XY Plot 1 m1 Curve Info Name m1 X 60.0000 60.0000 Y -0.4543 -27.6551 CPWA2CPS1 -5.00 dB(S(LumpPort1,LumpPort1)) Setup1 : Sw eep1 dB(S(LumpPort1,LumpPort2)) Setup1 : Sw eep1 -10.00 m2 Y1 -15.00 -20.00 -25.00 m2 -30.00 57.00 Freq [GHz] 62.00 Lecture 27 summary • Folded waveguide with thicker latch plates – Increase loss to 15.8 dB for 50 um thickness – Increase loss to 17.7 dB for 100 um thickness – Both structures are to be fabricated and tested • The transition of FWG to Radiation achieved – S12 = ‐0.45 dB – S11 = ‐27.7 dB • Roughness plays important roles, but is not considered in this report Report 1 Lecture 27 16 Via‐less transformer/coupler • Multi‐conductor transmission line modal analysis 123 • Geometric symmetry – One‐by‐one mode – Transmission lines N=3 – Telegraphers’ Eq PMC Term 2 Term 1 Term 3 GND CPW 200/200/200 (um) where matrices EEE 591/445 Z R j L Lecture 28 Y G jC 100um G2 100um Si Si G1 S2 S1 Lecture 27 G2 G1 17 where and V(z) and I(z) are voltage and current on each 2 conductor in vector form. ̅ 2 Eigen equation could be written as 2 ̅ 2 A linear transformation to diagonalize the propagation matrixes YZ and ZY where T is transformation matrix, made by the eigen vectors. The modal TL equations are EEE 591/445 Lecture 28 Lecture 27 18 modal and line voltages (current) are related by Vˆ T 1 V, Iˆ W 1 I From HFSS simulation, we extract from the geometry / vp Clearly, the quasi‐TEM Eqs (7‐69)‐(7‐71) are incorrect v (pe ) v (p0 ) v p For mode 1,2,3 0.02 V 3 EEE 591/445 Lecture 28 Lecture 27 19 For mode 2, the factor in term 3 is close to 0 and difference between term 1 and term 2 is just 4%. This mode like a coaxial mode, as shown 1 2 3 EEE 591/445 Lecture 28 Lecture 27 20 Coupler design • The coupler design is based on two coupled transmission lines • On Port 1 end, line 2 cannot be shorted to ground due to Port 2 Port 2 technical restriction. • We can only short line 2 to line 3, as 12 123 shown in the Fig. • Mode 2 is no use. Port 1 Port 1 • From previous modal analysis, this topology will short both port 1 and port 2 for mode 2. So mode 1 and mode 3 are the working modes in the coupler. The average quarter wavelength is av 2 413 μm. 4 4( 1 3 ) / 2 Lecture 28 123 EEE 591/445 Lecture 27 21 Final design A-A’ MEMS side CPW Cross sectional view of 200/200/200(um) quarter-wavelength 100um coupling section. 100um silicon silicon Back side CPW (um) A Back side layout 410um A’ MEMS Side Gold Silicon Back Side Gold Air Ai MEMS side side layout CPW Port 2 Port CPW Port 1 Port um Port 2 Port 1 EEE 591/445 Lecture 28 Lecture 27 22 • Transition loss is lower than 0.4 dB in 57GHz ‐ 64 GHz band. • Coupler to comb‐line filter – Whole structure 4dB loss. Ansoft LLC Name m10.00 62.2000 -3.6236 m2 64.0000 -4.0225 -5.00 -10.00 -15.00 X Y XY Plot 5 m1 m2 ANSOFT Curve Info dB(S(WavePort1,WavePort2)) Setup1 : Sweep1 h_beam='1.05um' dB(S(WavePort1,WavePort2)) Setup1 : Sweep1 h_beam='2.2um' dB(S(WavePort1,WavePort1)) Setup1 : Sweep1 h_beam='1.05um' dB(S(WavePort1,WavePort1)) Setup1 : Sweep1 h_beam='2.2um' 1mm Y1 -20.00 -25.00 -30.00 -35.00 -40.00 40.00 Back input ports input ports 50.00 60.00 Freq [GHz] 70.00 80.00 EEE 591/445 Lecture 28 Lecture 27 23 Collimated Antenna System Schematic system diagram system diagram 89 GHz Lumped component implementation component implementation FSS filter 54GHz Courtesy BUAA, 2008 Lecture 27 24 Collimated • Three elliptic reflectors truncated each contains 99.97% energy of dominating mode. – Reflection loss 0.005 dB. • Two channels of 54GHz and 89GHz share a common compact range system. • Gaussian beam source – corrugated horns – The ‐8.68dB Beam‐width 200 • Total loss in each channel 1 dB • Cross‐channel coupling 20 dB D 4 w0 Lecture 27 25 Collimated antenna: frequency selective surface (FSS) Component Courtesy BUAA, 2008 Lecture 27 26 Daubechies Discrete Wavelets of Frames: Diffraction Gaussian Beam Approach (DGBA) • Diffraction is treated by considering the edge at the nearest rim point to be straight. • Those beams that hit the reflector at less than two wavelength distance from the rim, suffer significant diffraction. • The elementary beams in Gaussian beam expansion are 12 wavelengths wide i.e. L=6λ. • The input plane is 52λ by 52λ with sampling shift λ, 0.8m away from the feed. The output plane is 2m away from the sub‐reflector. Lecture 27 27 Modeling by DGBA Lecture 27 28 Simulation Results Our simulation results Results from reference paper Lecture 27 29 Electronic packaging • Electromagnetic effects are described by Maxwell’s Equations in terms of field quantities Ex , Ey , Ez and Hx , Hy , Hz • Design engineers prefer distributed circuit arameters C, L, R, G (in matrix form) • Circuit Parameters were from low frequency models: – – – – C — Capacitance matrix, electric property of system, ε. L — Inductance matrix, magnetic property of system, μ. R — Resistance matrix, conductor loss, σc , R ∝√f. G — Conductance matrix, dielectric leakage, σd. Lecture 27 Quasi Quasi‐TEM ‐ 2 • At high frequencies, C, L, R, G become frequency dependent and must be frequency dependent and must be redefined. – C almost independent of frequency – L = μ0 ε0[C0]‐1 is no longer valid! – R ∝√f will result in Rdc=0, will incorrect! – G = ‐Im[Ĉ] is not valid! Lecture 27 Quasi Quasi‐TEM ‐ 3 Capacitances of the two line system of the two line system are almost frequency independent Inductances vary by frequency, particularly at low end Lecture 27 Quasi Quasi‐TEM ‐ 4 Resistance (left) and Conductance (right) vary strongly with frequency Lecture 27 General Configuration of Multiconductor Multilayer Transmission Lines (MMTL) Lecture 27 Quasi Quasi‐Dynamic Modeling of Skin‐Effect Resistance and Total Inductance • Combines full‐wave model of conductor interior with quasi‐static model between conductors • Quasi‐dynamic model dramatically improves computation time and accuracy – Printed circuit board microstrip under 10 GHz modeled quasi‐statically • 50% overestimate for resistance. • 30% underestimate for inductance. Lecture 27 Quasi‐Dynamic Formulation ‐ 1 – Inside the conductors (2 j ) J z 0 – Outside the conductors 2 Az 0 – Boundary conditions J z j n J z j l Az n Az l J z j [ Az Aq ] Lecture 27 Coiflet Implementation of Method of Moments ‐ 1 J z jm Bm (l ) m J z km Bm (l ) n m • Matrix equation V0 W0 U 0 K 0 0 j A0 j I S 0 0 Vi 0 U i J Lecture 27 Circuit Parameter Extraction J z * all wires dl ' Im{J z n } R L j 2 J z signal wire dl n Rij Lij J z * all wires dl Re{ Aq n } 2 J z signal wire dl n 1 P ( Rii R jj 2 d ) 2 2 Ix 1 Wm ( Lii L jj 4 2 ) 2 Ix • The inductance value computed from the quasi‐ static analysis can produce errors exceeding 300% Lecture 27 Wavelet Implementation of Microstrip Line Simulation ‐ 1 ( x ) 2 hk (2 x k ) • Wavelet‐sparsified impedance matrix is employed to study microstrip line problem 4 7 Three rectangular conductors over a ground plane Lecture 27 Wavelet Wavelet Implementation ‐ 2 Self and mutual resistances and inductances for three rectangular wires over a ground and mutual resistances and inductances for three rectangular wires over ground plane Lecture 27 Magnetic‐Head Connector (MHC) Example ‐ 1 – Configuration of high speed magnetic head connector. – Simulation determines the current distribution in ground plane and in the two circular wires at 1 GHz Lecture 27 Magnetic Magnetic Head Connector ‐ 2 Current distribution in the ground plane Lecture 27 Magnetic Magnetic Head Connector ‐ 3 Fig. 6: Current distribution in the left and right circular wire Lecture 27 Measurement and simulation Results for MHC – Measurements from 100 KHz to 1 GHz Fig. 9: Self and mutual resistances, R11, R12 by different methods Lecture 27 Measurement and Simulation Results Fig. 10: Self and mutual resistances, L11, L12 by different methods Quasi-static error may exceed 300%!! error may exceed 300%!! Lecture 27 General Edge Element Approach C 1 L ( Y11Y22 ) Y12 • Numerical examples Lecture 27 General Edge Element Approach A square spiral inductor as modeled by the finite-edge element method Scattering parameters Lecture 27 Rationale ● Problem - Traditional transmission lines at high frequencies lead to prohibitive li hi attenuation in a PCB channel ● Focus of research - Investigate hollow rectangular waveguides integrated into PCB to overcome the low pass characteristic of transmission lines ● Scope – W-band waveguides (70-110 GHz). • Dimensions allow for integration in a PCB Di PCB 0 Losses (dB / Inch) Insertion Loss (dB / Inch ) -1 -2 -3 -4 -5 -6 -7 -8 -9 W -band waveguide microstrip -10 0 10 20 30 40 50 60 70 80 90 100 110 Freguency (GHz) Lecture 27 48 A term PCBW means Printed Circuit Board integrated Waveguide An example of a PCBW: a hollow rectangular waveguide for W band ( 70~110 GHz ) integrated in a PC board 100 x 50 mils PCBW cross section from a test board Board material Hollow Metal walls Lecture 27 49 Manufacturing non-idealities of a real PCBW 100 x 50 mils PCBW cross section from a test board Glue protruding into waveguide Grooves affecting a current flow f low deformation of walls of walls • Sensitivity of the non-idealities on waveguide performance must be comprehended Lecture 27 50 Models of the PCBW manufacturing non-idealities PCBW Grooves are characterized by: ● width (from design) and height (less controllable, from manufacturing tech.); ● dielectric constant and loss tangent of their filling (glue, resin) ● Grooves are of the major impact on the field distribution, BW and attenuation glue h w Lecture 27 51 Configurations of PCB waveguides analyzed The difference is from manufacturing and in the number & location of grooves 1A 1B 1C 2 groove ● have the lowest attenuation and widest pass-band 30 100x50 PCBWs 30 100x50 PCBWs 30 100x50 PCBWs 30 100x50 PCBWs 25 25 25 25 A tte n u a t io n (d B /m ) A tte n u a t io n (d B /m ) 20 20 20 A tte n u a tio n (d B /m ) 60 70 80 90 100 110 120 130 A tte n u a tio n (d B /m ) 20 15 15 15 15 10 10 10 10 5 5 5 5 0 60 70 80 90 100 110 120 130 0 60 70 80 90 100 110 120 130 0 0 60 70 80 90 100 110 120 130 frequency (GHz) (GHz) frequency (GHz) (GHz) frequency (GHz) (GHz) frequency (GHz) (GHz) geom. ideal TE 10 h=2 mils h=1 mils h=0.5 mils Groove width = 17 mils Lecture 27 52 PCBW performance Attenuation (dB/m) 30 100x50 PCBW s 25 60 Useable BW of 100 x 50 mils PCBWs B W o f 100 50 m ils P CBWs 20 15 Useable BW (5dB) Useable bandwidth (GHz) 1A 50 10 5 40 0 60 70 80 90 100 110 120 130 30 frequency (GHz) 1C No grooves PCBW 1A, h = 2 mils PCBW 1A, h = 1 mils 4 6 8 10 12 14 16 18 20 22 24 20 10 0 P CBW 1A , h = 0. 5 m ils PCBW 1C h = 0.5~2 mils Groove width (mils) ● Useable BW / attenuation is affected by groove height and width ● For the best performance: th 12 < groove width < 17 mils; groove height – small as possible; loss tangent of filling – small as possible ● PCBW 1C exhibits almost ideal behavior Lecture 27 53 Launch structures Transition with probe via Transition with probe ( via antenna ) waveguide port probe (via antenna) port Finline transition from / to microstrip waveguide port microstrip port Lecture 27 54 Transition with probe ( via antenna ) Port Via (antenna) PCB dielectric “Open” Boundary Conditions Resin in the groove Lecture 27 55 Fin-line transition from microstrip to PCB waveguide PCBW 500 ohm at 90 GHz at 90 GHz waveguide port antipodal finline microstrip 50 ohm microstrip port ● Dielectric slab is to support the finline, needed only for the length of the transition ● E-field makes rotation on 90 degrees in the transition & gets its longitudinal component Lecture 27 56 How it was simulated Transition, launch structure [S0] Full - wave simulation with HFSS ith HFSS PCPW channel with launchers: [Ts] = [T1][T2][T3], [Ss] [S1], [T1] [S2], [T2] [S2], [T3] Cascaded ● It allows fast evaluating PCBW sections of different lengths without loss of accuracy Lecture 27 57 Performance of the launch structures Probe port 0 PCBW po port μstrip port PCBW po port 1 00 x 5 0 m ils P C B W transition , h = 2 m ils, w = 1 6 m ils 0 100 x 50 mils strip-PCBW transition insertion/reflection loss (dB) insertion/reflection loss (dB) -5 -5 S21 S12 S11 S22 -10 S 11 -15 S 22 S 21 -20 S 12 -10 -15 metrics -20 -25 -25 -30 60 70 80 90 100 110 -30 60 70 80 90 100 110 Frequency (GH z) 3 Frequency (GHz) 3 2 .8 via port P CB W port 1 0 0 x 5 0 m ils P C B W trans itio n, h = 2 m ils , w = 1 6 m ils 1 0 0 x 5 0 m ils s trip -P C B W tra ns itio n V S W R 2.8 2.6 2.4 2 .6 2 .4 VSWR VSWR 2.2 2 1.8 1.6 1.4 1.2 1 60 65 70 75 80 85 90 95 100 105 110 2 .2 2 1 .8 1 .6 1 .4 1 .2 1 60 65 70 75 80 85 90 s trip port P C B W port 95 100 1 05 110 Frequency (G H z) F re q ue nc y (G H z ) ● Finline launch structure outperforms that with via antenna by ~ 12 dB in |S11| Lecture 27 58 Channel response 1 0 0 x 5 0 m ils P C B W 1 A s e c tio n, le ng th = 3 " 0 0 1 00 x 50 mils PCBW section, length = 3" -5 Insertion / Reflection loss (dB) Insertion / Reflection loss (dB) -5 -1 0 -10 -1 5 S 21 S 12 S 11 S 22 -15 -2 0 -20 -2 5 3” 60 70 80 90 100 110 -25 -3 0 matched PCBW - strip matched PCBW S21 S12 S11 S22 60 70 80 90 100 3” 110 -30 F re q ue nc y (G H z ) 0 Frequency (GHz) 0 1 0 0 x 5 0 m ils P C B W 1 A s e c tio n, le ng th = 1 0 " 1 00 x 50 mils PCBW section, length = 10" -5 Insertion / Reflection loss (dB) Insertion / Reflection loss (dB) -5 matched PCBW - strip matched PCBW S21 S12 S11 S22 -10 -10 -15 S 21 S 12 S 11 S 22 -15 -20 -20 -25 10” 60 70 80 90 100 1 10 -25 10” 60 70 80 90 100 110 -30 -30 F re q ue nc y (G H z ) Frequency (GHz) ● PCBW sections with finline launchers outperform ones with vias in pass band flatness and return loss Lecture 27 59 What do we have for channels (3”~10”) (3” with PCBW - microstrip finline transition in in W-band ● Return loss level below -10 dB: in 70 - 110 GHz loss level below dB: in 70 GHz ● Return loss level below -15 dB: in 80 - 105 GHz ● Pass-band rippleness: smaller than 0.8 dB at 0.4 GHz ● Transmission loss: 3 dB – 5 dB in 70 - 110 GHz Lecture 27 60 ...
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