MIT6_012F09_lec11

MIT6_012F09_lec11 - 6.012 - Microelectronic Devices and...

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Unformatted text preview: 6.012 - Microelectronic Devices and Circuits Lecture 11 - MOSFETs II; Large Signal Models - Outline • Announcements On Stellar - 2 write-ups on MOSFET models • The Gradual Channel Approximation (review and more) MOSFET model: gradual channel approximation iD ≈ 0 K K (Example: n-MOS) for (vGS – VT)/α ≤ 0 ≤ vDS (cutoff) 2 /2α (vGS – VT) for 0 ≤ (vGS – VT)/α ≤ vDS (saturation) (vGS – VT – αvDS/2)vDS for 0 ≤ vDS ≤ (vGS – VT)/α (linear) with K ≡ (W/L)µeCox*, VT = VFB – 2φp-Si + [2εSi qNA(|2φp-Si| – vBS)]1/2/Cox* and α = 1 + [(εSi qNA/2(|2φp-Si| – vBS)]1/2 /Cox (frequently α ≈ 1) • Refined device models for transistors (MOS and BJT) Other flavors of MOSFETS: p-channel, depletion mode The Early Effect: 1. Base-width modulation in BJTs: wB(vCE) 2. Channel-length modulation in MOSFETs: L(vDS) Charge stores: 1. Junction diodes 2. BJTs 3. MOSFETs Extrinsic parasitics: Lead resistances, capacitances, and inductances Clif Fonstad, 3/18/08 Lecture 11 - Slide 1 An n-channel MOSFET showing gradual channel axes S G + vGS – n+ 0 0 iG D iD n+ x p-Si vBS vDS + B iB L y Extent into plane = W Gradual Channel Approximation: - A one-dimensional electrostatics problem in the x direction is solved to find the channel charge, qN*(y); this charge depends on vGS, vCS(y) and vBS. - A one-dimensional drift problem in the y direction then gives the channel current, iD, as a function of vGS, vDS, and vBS. Clif Fonstad, 3/18/08 Lecture 11 - Slide 2 Gradual Channel Approximation i-v Modeling (n-channel MOS used as the example) The Gradual Channel Approximation is the approach typically used to model the drain current in field effect transistors.* It assumes that the drain current, iD, consists entirely of carriers flowing in the channel of the device, and is thus proportional to the sheet density of carriers at any point and their net average velocity. It is not a function of y, but its components in general are: iD = " W # "q # n * ( y ) # sey ( y ) ch In this expression, W is the width of the device, -q is the charge on each electron, n*ch(y) is sheet electron concentration in the channel (i.e. electrons/cm2) at y, and sey(y) is the net ! electron velocity in the y-direction. If the electric field is not too large, sey(y) = -µeEy(y), and iD = " W # q # n * ( y ) # µe E y ( y ) = W # q # n * ( y ) # µe ch ch dvCS ( y ) dy Cont. * Junction FETs (JFETs), MEtal Semiconductor FETs (MESFETs1), and Heterojunction FETs 2 Clif Fonstad, 3/18/08 (HJFETs ), as well as Metal Oxide Semiconductor FETs (MOSFETs). Lecture 11 - Slide 3 1. Also called Shottky Barrrier FETs (SBFETs). 2. Includes HEMTs, TEGFETs, MODFETs, SDFETs, HFETs, PHEMTs, MHEMTs, etc. ! GCA i-v Modeling, cont. S – n+ We have: G + vGS 0 0 dvCS ( y ) iD = W " q " n ( y ) " µe dy vDS iG vBS iD n+ x L p-Si * ch D + B y iB To eliminate the derivative from this equation we integrate both sides with respect to y from the source (y = 0) to the drain (y = L). This corresponds to integrating the right hand side with respect to vCS from 0 to vDS, because vCS(0) = 0 to vCS(L) = vDS: L L dvCS ( y ) * " iD dy = W # µe # q # " nch ( y ) dy dy = W # µe # q # 0 0 v DS n * (vCS ) dvCS " ch 0 The left hand integral is easy to evaluate; it is simply iDL. Thus we have: L W " iD dy = iD L # iD = L $ µe $ q $ 0 Clif Fonstad, 3/18/08 v DS n * (vCS ) dvCS " ch 0 Cont. Lecture 11 - Slide 4 GCA i-v Modeling, cont. The various FETs differ primarily in the nature of their channels and thereby, the expressions for n*ch(y). For a MOSFET we speak in terms of the inversion layer charge, qn*(y), which is equivalent to - q·n*ch(y). Thus we have: v DS W iD = " µe # q* (vGS , vCS , v BS ) dvCS n L 0 We derived qn* earlier by solving the vertical electrostatics problem, and found: * q* (vGS , vCS , v BS ) = " Cox [vGS " vCS " VT (vCS , v BS )] n ! { [ with VT (vCS , v BS ) = VFB " 2# p " Si + 2$SiqN A 2# p " Si " v BS + vCS } 1/ 2 * Cox Using this in the equation for iD, we obtain: ! W iD (vGS , v DS , v BS ) = µe L v DS # {C [v * ox GS " vCS " VT (vCS , v BS )]} dvCS 0 At this point we can do the integral, but it is common to simplify the expression of VT(vCS,vBS) first, to get a more useful result. ! Clif Fonstad, 3/18/08 Cont. Lecture 11 - Slide 5 GCA - dealing with the non-linear dependence of VT on vCS Approach #1 - Live with it Even though VT(vCS,vBS) is a non-linear function of vCS, we can still put it in this last equation for iD: W iD = µe L + *% t ,Cox 'vGS " vCS " VFB + 2# p " Si " ox 2$SiqN A 2# p " Si " v BS + vCS 1& $ox 0- v DS [ (. */ dvCS )0 and do the integral, obtaining: * W v' *$ iD (v DS , vGS , v BS ) = µe Cox +& vGS " 2# p " VFB " DS )v DS L 2( ,% ( 3 . + 2-SiqN A 0 2# p + v DS " v BS / 2 ! ) 3/2 ( " 2# p " v BS ) 3/2 14 35 26 The problem is that this result is very unwieldy, and difficult to work with. More to the point, we don't have to live with it because it is easy to get very good, approximate solutions that are much simpler to work with. Clif Fonstad, 3/18/08 Cont. Lecture 11 - Slide 6 GCA - dealing with the non-linear dependence of VT on vCS Approach #2 - Ignore it Early on researchers noticed that the difference between VT at 0 and at y, i.e. VT(0,vBS) and VT(vDS,vBS), is small, and that using VT(0,vBS) alone gives a result that is still quite accurate and is very easy to use: v DS W * iD (vGS , v DS , v BS ) = µe # {Cox [vGS " vCS " VT (0, v BS )]} dvCS L 0 2 $ W v DS ' * = µe Cox %[vGS " VT (0, v BS )] v DS " ( L 2) & The variable, vCS, is set to 0 in VT. This result looks much simpler than the result of Approach #1, and it is much easier to use in hand calculations. It is, in ! fact, the one most commonly used by the vast majority of engineers. At the same time, the fact that it was obtained by ignoring the dependence of VT on vCS is cause for concern, unless we have a way to judge the validity of our approximation. We can get the necessary metric through Approach #3. Clif Fonstad, 3/18/08 Cont. Lecture 11 - Slide 7 GCA - dealing with the non-linear dependence of VT on vCS Approach #3 - Linearize it (i.e. expand it, keep first order term) In this approach we leave the variation of VT with vCS in, but linearize it by doing a Taylor's series expansion about vCS = 0: #VT VT [vCS , v BS ] " VT (0, v BS ) + $ vCS #vCS v = 0 CS Taking the derivative and evaluating it at vCS = 0 yields: t ox VT [vCS , v BS ] " VT (0, v BS ) + ! #ox #SiqN A & vCS 2 2$ p % v BS ( ) With this qn* is ! ' t * q* (vGS , vCS , v BS ) " # Cox )vGS # vCS + VT (0, v BS ) # ox n ) $ox ( $SiqN A 2 2% p # v BS ( ) * & vCS , , + * = # Cox [vGS # - vCS + VT (v BS )] where ! Clif Fonstad, 3/18/08 ' t ox * " # )1 + $SiqN A 2 2% p & v BS , and VT (v BS ) # VT (0, v BS ) ( $ox + Lecture 11 - Slide 8 ( ) Cont. GCA - dealing with the non-linear dependence of VT on vCS Using this result in the integral in the expression for iD gives: v DS W * iD (vGS , v DS , v BS ) = µe $ {Cox [vGS " # vCS " VT (0, v BS )]} dvCS L 0 2 % W v DS ( * = µe Cox &[vGS " VT (v BS )] v DS " # ) L 2* ' Except for α this is the Approach 2 result. Plotting this equation for increasing values of vGS we see that it traces inverted parabolas as shown below. ! iD inc. vGS Note: iD saturates after its peak value (solid lines), rather than decreasing (dashed lines). vDS Clif Fonstad, 3/18/08 Cont. Lecture 11 - Slide 9 Gradual Channel Approximation, cont. The drain current expression, cont: The point at which iD reaches its peak value and saturates is easily found. Taking the derivative and setting it equal to 1 zero we find: " iD =0 when v DS = [vGS $ VT (v BS )] " v DS # What happens physically at this voltage is that the channel (inversion) at the drain end of the channel disappears: ! * q* ( L) " # Cox {vGS # VT (v BS ) # $ v DS } n 1 = 0 when v DS = [vGS # VT (v BS )] $ For vDS > [vGS-VT(vBS)]/α, all the additional drain-to-source voltage appears across the high resistance region at the drain end of ! the channel where the mobile charge density is very small, and iD remains constant independent of vDS: iD (vGS , v DS , v BS ) = Clif Fonstad, 3/18/08 ! 1W 2 * µe Cox [vGS # VT (v BS )] for 2" L v DS > 1 [vGS # VT (v BS )] " Lecture 11 - Slide 10 Gradual Channel Approximation, cont. The full model: With this drain current expression, we now have the complete set of Gradual Channel Model expressions for the MOSFET terminal characteristics in the three regions of operation: Valid for v BS " 0, and v DS # 0 : iG (vGS , v DS , v BS ) = 0 and iB (vGS , v DS , v BS ) = & , 0 , , 1W 2 * iD (vGS , v DS , v BS ) = ' µe Cox [vGS $ VT (v BS )] 2%L , v) ,W *& µe Cox 'vGS $ VT (v BS ) $ % DS * % v DS , %L ( 2+ ( with VT (v BS ) - VFB Clif Fonstad, 3/18/08 { [ 0 for [vGS $ VT (v BS )] < 0 < % v DS 0 < [vGS $ VT (v BS )] < % v DS for 0 < % v DS < [vGS $ VT (v BS )] for 1 $ 2. p $ Si + * 2/SiqN A 2. p $ Si $ v BS Cox & 1, /SiqN A % - 1+ * ' Cox , 2 2. p $ Si $ v BS ( [ )1 / 2 , * , + } C* ox 1/ 2 /ox t ox Lecture 11 - Slide 11 iD Gradual Channel Approximation, cont. ! $ * 0 * * K 2 iD (vGS , v DS , v BS ) =! % [vGS " VT (v BS )] 2 * * K $v " V (v ) " # v DS ' # v ( DS T BS * % GS & 2) & Linear or Triode Region Clif Fonstad, 3/18/08 iD iB G+ iB (vGS , v DS , v BS ) = 0 iG (vGS , v DS , v BS ) = 0 + vDS iG The full model, cont: D + B vBS vGS – for S [vGS " VT (v BS )] < 0 < # v DS for 0 < [vGS " VT (v BS )] < # v DS Saturation for 0 < # v DS < [vGS " VT (v BS )] Linear or Triode increasing with vGS K+ Cutoff W * µe Cox #L Saturation or Forward Active Region Cutoff Region vDS Lecture 11 - Slide 12 The operating regions of MOSFETs and BJTs: Comparing an n-channel MOSFET and an npn BJT iD MOSFET D + Linear iD or Triode iG Saturation (FAR) vDS G+ iD ! K [vGS - V T(vBS)]2/2! vGS – iC BJT C – S + Cutoff vDS iB vCE B+ Saturation iC i vBE – B – E FAR iB ! IBSe qV BE /kT vCE > 0.2 V Cutoff 0.6 V Input curve Clif Fonstad, 3/18/08 vBE Forward Active Region iC ! !F iB 0.2 V Cutoff vCE Output family Lecture 11 - Slide 13 p-channel MOSFET's: The other "flavor" of MOSFET p-channel S – vGS G + iG vDS p+ D iD p+ Structure: n-Si vBS The voltage progression: Inversion + B Depletion VT Gradual channel model*: iB Accumulation v GS 0 V FB For enhancement mode p-channel: VT ( i.e. v GS at threshold) < 0, VFB > 0.. Valid for v SB " 0, and v SD # 0 : iG (v SG , v SD , v SB ) = 0 and iB (v SG , v SD , v SB ) = 0 & , 0 for [v SG $ VT (v SB ) ] < 0 < % v SD , , 2 1W * $ iD (v SG , v SD , v SB ) = ' µe Cox [v SG $ VT (v SB ) ] for 0 < [v SG $ VT (v SB ) ] < % v SD 2%L , v) ,W *& µe Cox 'v SG $ VT (v SB ) $ % SD * % v SD for 0 < % v SD < [v SG $ VT (v SB ) ] , %L ( 2+ ( 1 1/ 2 VT (v SB ) = VFB " 2# n " Si " $ [2# n " Si " v SB ] with $ % * [2&SiqN D ] 1 / 2 Cox Clif Fonstad, 3/18/08 Lecture 11 - Slide 14 ! * Enhancement mode only, VT ( i.e. v GS at threshold) < 0. p-channel MOSFET's: cont. S G + vGS – vDS iG p+ p-channel Structure: n-Si + Symbol: vDS iG iB G+ + vBS B FAR model: vGS < VT iG vBS > 0 vDS < 0 G + – – + Symbol: vSG vSB – – B iB iG vSD -iD – D FAR model: vSG > -VT vSB < 0 vSD > 0 + iB vDS i D(vGS ,vDS,vBS ) B iB + vBS vGS S Oriented as found in circuits: S + + B D iD vBS vGS Clif Fonstad, 3/18/08 iD p+ Symbol and FAR model: Oriented with source down like n-channel: iD D G D – S S + + vSB iB vSG – – B -i D(vSG ,vSD,vSB ) vSD G iG -iD – D Lecture 11 - Slide 15 Depletion mode MOSFET's: The very last MOSFET variant It is possible to have n-channel MOSFETs with VT < 0. In this situation the channel exists with vGS = 0, and a negative bias must be applied to turn it off. This type of device is called a "depletion mode" MOSFET. Devices with VT > 0 are "enhancement mode." iD increasing vGS vGS = 0 vGS ! V T vDS For a p-channel depletion mode MOSFET, VT > 0. The expressions for iD(vGS, vDS, vBS) are exactly the same for enhancement mode and depletion mode MOSFETs. Clif Fonstad, 3/18/08 Lecture 11 - Slide 16 BJT Characteristics (npn) Saturation iC iB FAR Forward Active Region iB ! IBS e qV BE /kT vCE > 0.2 V iC ! !F iB Cutoff vBE 0.6 V 0.2 V Input curve BJT MODELS BJT Models Output family –– BC vBC IIES ES vBE vBE iFF i "!iRiR RR – Clif Fonstad, 3/18/08 CC + ICS ICS iRiR + "FiiF !F F B+ B+ vCE Cutoff – E – E Forward active region: vBE > 0.6 V vCE > 0.2 V CC !F!bFiB i (i.e. vBC < 0.4V) iR is negligible vCE Other regions Cutoff: vBE < 0.6 V Saturation: vCE < 0.2 V i iB B B+ B + IBS IBS vvBE BE – – EE Lecture 11 - Slide 17 MOSFET Characteristics Linear iD or Triode Saturation (FAR) (n-channel) iD ! K [vGS - V T(vBS )]2/2! Also: iG ≈ 0 iB ≈ 0 K = (W/L)µeCox* Cutoff vDS Output family Output family Model valid for vBS ≤ 0 and vDS ≥ 0, insuring iG(vGS, vDS,vBS) ≈ 0, iB(vGS, vDS,vBS) ≈ 0 0 MOSFET D circuit model iD + vDS iG G + vGS i D(vGS ,vDS,vBS ) ≈ iB + B vBS –– S Clif Fonstad, 3/18/08 for (vGS – VT) ≤ 0 ≤ αvDS (cutoff) (W/2αL)µeCox*(vGS – VT)2 for 0 ≤ (vGS – VT) ≤ αvDS (saturation) (W/αL)µeCox*(vGS – VT – αvDS/2)αvDS for 0 ≤ αvDS ≤ (vGS – VT) (linear) with VT = VFB – 2φp-Si + [2εSi qNA(|2φp-Si| – vBS)]1/2/Cox* α = 1 + [(εSi qNA/2(|2φp-Si| – vBS)]1/2 /Cox (frequently α ≈ 1) Lecture 11 - Slide 18 The Early Effect: (exaggerated for clarity*) BJT: iC iC npn 0.2 V -1/! = -V A MOSFET: iD vvCE CE iD n-channel -1/! = -V A Clif Fonstad, 3/18/08 * Typically the Early effect is far more important in small-signal applications than large signal. vDS v DS Lecture 11 - Slide 19 Active Length Modulation - the Early Effect: MOSFET "Channel length modulation" MOSFET: We begin by recognizing that the channel length decreases with increasing vDS and writing this dependence to first order in vDS: 1 [1 + $(v DS # VDSat )] L " Lo [1 # $(v DS # VDSat )] and " L Lo W * K= µe Cox %L Inserting the channel length variation with vDS into K we have: ! K " K o [1 + #(v DS $ VDSat )] where Ko % W * µe Cox & Lo Thus, in saturation: iD " Ko 2 (vGS # VT ) [1 + $(v DS # VDSat )] 2 ! Note: λ is the inverse of the Early Voltage, VA (i.e., λ = 1/VA). Clif Fonstad, 3/18/08 Lecture 11 - Slide 20 ! Active Length Modulation - the Early Effect: BJT "Base width modulation" BJT: We begin by recognizing that the base width decreases with increasing vCE and writing this dependence to first order in vCE: 1 1 w * " w * (1 # $vCE ) and " * (1 + $vCE ) B Bo w * w Bo B Then we recall that in a modern BJT the base defect, δB, is negligible and βF depends primarily on the emitter defect, δE, and can be written: ! (1 + #B ) $ 1 = De N DE w * E "F = (#E + #B ) #E Dh N AB w * B Inserting the base width variation with vCE into βF we have: * De N DE wE "F ! "Fo(1 + $vCE ) where "Fo % # * Dh N AB wBo Thus, in the F.A.R.: iC " # Fo (1 + $vCE ) iB Note: λ is the inverse of the Early Voltage, VA (i.e., λ = 1/VA). ! Clif Fonstad, 3/18/08 Lecture 11 ! - Slide 21 Large signal models*: i BJT: npn Saturation iC FAR Forward Active Region iC ! !F iB iB ! IBSe qV BE /kT vCE > 0.2 V Cutoff iC ≈ βF(1 + λvCE)iB 0.6 V MOSFET: n-channel vBE Linear iD or Triode iD ≈ K[vGS - VT(vBS) - vDS/2]vDS 0.2 V Saturation (FAR) ! v [vGS - V T(vBS 2 2/2! iD ≈iDK[K GS-VT(vBS)])][1+λ(vDS-VDSat]/2 Cutoff Clif Fonstad, 3/18/08 vCE Cutoff vDS * The Early effect is included, but barely visible. Lecture 11 - Slide 22 Large signal models: when will we use them? Digital circuit analysis/design: This requires use of the entire circuit, and will be the topic of Lectures 14, 15, and 16. Bias point analysis/design: This uses the FAR models (below and Lec. 17ff). C C iIC = β IB C BJT !FiB IB iB iB B+ B+ IBS 0.6 vBE V vBE – – E E D MOSFET D i D = (K/2)[vGS - V T] G G VGS = (2IDS/K)1/2 + vVS T vGS – Clif Fonstad, 3/18/08 2 G+ + S iID D + B vBS – S Lecture 11 - Slide 23 Charge stores in devices: we must add them to our device models Parallel plate capacitor qA " qA ,PP = A v AB d ρ(x) d/2 $qA ,PP C pp (VAB ) # $v AB x -d/2 qB( = -qA) Depletion region charge store ρ(x) qNDn v AB = VAB qA ,DP (v AB ) = " A 2q#Si [$ b " v AB ] ! qB -xp A" d = N Ap N Dn [N Ap + N Dn ] ( = -Q A ) x xn qA Cdp (VAB ) = A -qNAp N Ap N Dn q#Si 2[$ b " VAB ] [ N Ap + N Dn ] QNR region diffusion charge store p’(x), n’(x) qAB ,DF (v AB ) " Aqn i2 p’(xn) ! qA, qB (=-qA) n’(-xp ) -wp -xp xn x Cdf (VAB ) " Clif Fonstad, 3/18/08 A #Si w (VAB ) Dh [e qVAB / kT # 1] N Dn w n ,eff Note: Approximate because we are only accounting for the charge store on the lightly doped side. wn = 2 w n ,eff q ID (VAB ) 2 Dh kT = 2 w n ,eff 2 Dh iD (v AB ) Lecture 11 - Slide 24 Adding charge stores to the large signal models: A p-n diode: qAB IBS qAB: Excess carriers on p-side plus excess carriers on n-side plus junction depletion charge. B qBC C BJT: npn (in F.A.R.) iB’ B MOSFET: n-channel qBE: Excess carriers in base plus E-B junction depletion charge qBC: C-B junction depletion charge IBS E qBE D iD S qG: Gate charge; a function of vGS, vDS, qDB qG G Clif Fonstad, 3/18/08 !FiB’ B and vBS. qDB: D-B junction depletion charge qSB: S-B junction depletion charge qSB Lecture 11 - Slide 25 6.012 - Microelectronic Devices and Circuits Lecture 11 - MOSFETS II; Large-Signal Models - Summary • Gradual channel approximation for FETs General approach MOSFETS in strong inversion 1. Ignore ariation of VT along channel v 2. Linearize variation of VT along channel: introduces α factor • Additional device model issues The Early Effect: 1. Base-width modulation in BJTs: wB(vCE) In the F.A.R.: iC ≈ βFo(1 + lvCE)iB 2. Channel-length modulation in MOSFETs: L(vDS) In saturation: iD ≈ Ko (vGS – VT)2 [1 + λ(vDS-VDSat)]/2α Charge stores: 1. Junction diodes - depletion and diffusion charge 2. BJTs - at EB junction: depletion and diffusion charge at CB junction: depletion charge (focus on FAR) 3. MOSFETs - between B and S, D: depletion charge of n+-p junctions between G and S, D, B: gate charge (the dominant store) in cut-off: Cgs ≈ Cgd ≈ 0; all is Cgb linear region: Cgs = Cgd = W L Cox*/2 in saturation region: Cgs = (2/3) W L Cox* Cgd = 0 (only parasitic overlap) Clif Fonstad, 3/18/08 Lecture 11 - Slide 26 MIT OpenCourseWare http://ocw.mit.edu 6.012 Microelectronic Devices and Circuits Fall 2009 For information about citing these materials or our Terms of Use, visit: http://ocw.mit.edu/terms. ...
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This note was uploaded on 11/07/2011 for the course COMPUTERSC 6.012 taught by Professor Charlesg.sodini during the Fall '09 term at MIT.

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